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ISL6531
Data Sheet August 11, 2005 FN9053.2
Dual 5V Synchronous Buck Pulse-Width Modulator (PWM) Controller for DDRAM Memory VDDQ and VTT Termination
The ISL6531 provides complete control and protection for dual DC-DC converters optimized for high-performance DDRAM memory applications. It is designed to drive low cost N-channel MOSFETs in synchronous-rectified buck topology to efficiently generate 2.5V VDDQ for powering DDRAM memory, VREF for DDRAM differential signalling, and VTT for signal termination. The ISL6531 integrates all of the control, output adjustment, monitoring and protection functions into a single package. The VDDQ output of the converter is maintained at 2.5V through an integrated precision voltage reference. The VREF output is precisely regulated to 1/2 the memory power supply, with a maximum tolerance of 1% over temperature and line voltage variations. VTT accurately tracks VREF. During V2_SD sleep mode, the VTT output is maintained by a low power window regulator. The ISL6531 provides simple, single feedback loop, voltagemode control with fast transient response for the VDDQ regulator. The VTT regulator features internal compensation that eases the design. It includes two phase-locked 300kHz triangle-wave oscillators which are displaced 90o to minimize interference between the two PWM regulators. The regulators feature error amplifiers with a 15MHz gainbandwidth product and 6V/s slew rate which enables high converter bandwidth for fast transient performance. The resulting PWM duty ratio ranges from 0% to 100%. The ISL6531 protects against overcurrent conditions by inhibiting PWM operation. The ISL6531 monitors the current in the VDDQ regulator by using the rDS(ON) of the upper MOSFET which eliminates the need for a current sensing resistor.
Features
* Provides VDDQ, VREF, and VTT voltages for one- and twochannel DDRAM memory systems * Excellent voltage regulation - VDDQ = 2.5V 2% over full operating range -- VREF = 1 V DDQ 1% over full operating range 2 - VTT = VREF 30mV * Supports `S3' sleep mode -- VTT is held at 1 V DDQ via a low power window 2 regulator to minimize wake-up time * Fast transient response - Full 0% to 100% duty ratio * Operates from +5V Input * VTT regulator internally compensated * Overcurrent fault monitor on VDD - Does not require extra current sensing element - Uses MOSFET's rDS(ON) * Drives inexpensive N-Channel MOSFETs * Small converter size - 300kHz fixed frequency oscillator * 24 Lead, SOIC or 32 Lead, 5mmx5mm QFN * Pb-Free Plus Anneal Available (RoHS Compliant)
Applications
* VDDQ, VTT, and VREF regulation for DDRAM memory systems - Main memory in AMD(R) AthlonTM and K8TM, Pentium(R) III, Pentium IV, Transmeta, PowerPCTM, AlphaPCTM, and UltraSparc(R) based computer systems * High-power tracking DC-DC regulators
Ordering Information
PART NUMBER ISL6531CB ISL6531CBZ (See Note) ISL6531CR TEMP RANGE(oC) 0 to 70 0 to 70 0 to 70 PACKAGE 24 Lead SOIC 24 Lead SOIC (Pb-free) PKG. DWG. # M24.3 M24.3
32 Lead 5x5 QFN L32.5x5
ISL6530/31EVAL1 Evaluation Board
Add "-T" suffix for tape and reel. NOTE: Intersil Pb-free plus anneal products employ special Pb-free material sets; molding compounds/die attach materials and 100% matte tin plate termination finish, which are RoHS compliant and compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright (c) Intersil Americas Inc. 2003, 2005. All Rights Reserved. All other trademarks mentioned are the property of their respective owners.
ISL6531 Pinouts
24 LEAD (SOIC) TOP VIEW
BOOT1 BOOT1 UGATE1 BOOT1 1 2 24 PGND1 23 LGATE1 22 PVCC1 21 OCSET/SD 20 V2_SD 19 PGOOD 18 N/C 17 SENSE2 16 N/C 15 VCC 14 LGATE2 13 PGND2 SENSE1 VREF_IN GNDA GNDA 5 6 7 8 9 PHASE2 10 BOOT2 11 BOOT2 12 UGATE2 13 PGND2 14 PGND2 15 LGATE2 16 VCC 20 N/C 19 SENSE2 18 N/C 17 VCC VREF FB1 COMP1 2 3 4 23 OCSET/SD 22 V2_SD 21 PGOOD PHASE 1 1
32 LEAD 5X5 (QFN) TOP VIEW
UGATE1 UGATE1 LGATE1 26 PGND1 PGND1 PVCC1 25 24 PVCC1
PHASE1 3 VREF 4 FB1 5 COMP1 6
32
31
30
29
28
27
SENSE1 7 VREF_IN 8 GNDA 9
PHASE2 10 BOOT2 11 UGATE2 12
2
ISL6531 Block Diagram
PGOOD OCSET/SD VCC
POWER-ON RESET (POR)
40A + + + + -
+ OVERCURRENT
SOFTSTART
BOOT1
X 0.85
X 1.15
X 0.85
X 1.15
UGATE1 PHASE1 ERROR AMP + PWM COMPARATOR + GATE INHIBIT CONTROL LOGIC PWM PVCC1
FB1 COMP1 SENSE1 VREF_IN
0.8V REFERENCE OSCILLATOR
LGATE1
PGND1 VREF + 90o Phase Shift
BOOT2 ERROR AMP Zf + Zc + PWM COMPARATOR PWM INHIBIT GATE CONTROL LOGIC VCC LGATE2 PHASE2
UGATE2
SENSE2
WINDOW REGULATOR
V2_SD
PGND2
GND
3
ISL6531 Typical Application
+5V PGOOD
ROCSET VCC PGOOD BOOT1 OCSET/SD RESET GNDA PHASE1 +5V PVCC1 SLEEP V2_SD ISL6531 LGATE1 Q2 UGATE1 CBOOT1 VDDQ LOUT1 COUT1 Q1 DBOOT1
VREF (.5xVDDQ)
VREF_IN
PGND1
VREF DBOOT2 COMP1 BOOT2 UGATE2 CBOOT2 PHASE2 LOUT2 FB1 LGATE2 PGND2 SENSE1 RFB1 SENSE2 Q4 COUT2 VTT Q3
FIGURE 1. TYPICAL APPLICATION FOR ISL6531
4
ISL6531
Absolute Maximum Ratings
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +7.0V Boot Voltage, VBOOTn - VPHASEn . . . . . . . . . . . . . . . . . . . . . . +7.0V Input, Output or I/O Voltage . . . . . . . . . . . . GND -0.3V to VCC +0.3V ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 2
Thermal Information
Thermal Resistance JA (oC/W) JC (oC/W) SOIC Package (Note 1) . . . . . . . . . . . . 65 N/A QFN Package (Note 2). . . . . . . . . . . . . 33 4 Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . 150oC Maximum Storage Temperature Range . . . . . . . . . . -65oC to 150oC Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . 300oC (SOIC - Lead tips only) For Recommended soldering conditions see Tech Brief TB389.
Operating Conditions
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5V 10% Ambient Temperature Range . . . . . . . . . . . . . . . . . . . . . 0oC to 70oC Junction Temperature Range . . . . . . . . . . . . . . . . . . . . 0oC to 125oC
CAUTION: Stresses above those listed in "Absolute Maximum Ratings" may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTE: 1. JA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details. 2. JA is measured in free air with the component mounted on a high effective thermal conductivity test board with "direct attach" features. JC, the "case temp" is measured at the center of the exposed metal pad on the package underside. See Tech Brief TB379.
Electrical Specifications
PARAMETER VCC SUPPLY CURRENT Nominal Supply Shutdown Supply POWER-ON RESET Rising VCC Threshold Falling VCC Threshold OSCILLATOR Free Running Frequency REFERENCES Reference Voltage (V2 Error Amp Reference) V1 Error Amp Reference Voltage Tolerance V1 Error Amp Reference ERROR AMPLIFIERS DC Gain Gain-Bandwidth Product Slew Rate WINDOW REGULATOR Load Current Output Voltage Error GATE DRIVERS
Recommended Operating Conditions with Vcc = 5V, unless otherwise noted. SYMBOL TEST CONDITIONS MIN TYP MAX UNITS
ICC
OCSET/SD=VCC; UGATE1, UGATE2, LGATE1, and LGATE2 Open OCSET/SD=0V
-
5 3
-
mA mA
VOCSET/SD=4.5V VOCSET/SD=4.5V
4.25 3.75
-
4.5 4.0
V V
VCC=5
275
300
325
kHz
VVREF
SENSE1=2.5V
49.5 -
50.0 0.8
50.5 2 -
%SENSE1
%
VREF
VCC=5
-
V
GBW SR COMP=10pF -
82 15 6
-
dB MHz V/s
V2_SD=VCC; 10mA load on V2 -
10 7
-
mA %
Upper Gate Source (UGATE1 and 2) Upper Gate Sink (UGATE1 and 2) Lower Gate Source (LGATE1 and 2) Lower Gate Sink (LGATE1 and 2) PROTECTION OCSET/SD Current Source OCSET/SD Disable Voltage
IUGATE IUGATE ILGATE ILGATE
VCC=5V, VUGATE=2.5V VUGATE-PHASE=2.5V VCC=5V, VLGATE=2.5V VLGATE=2.5V
-
-1 1 -1 2
-
A A A A
IOCSET VRESET
VOCSET=4.5VDC
34 -
40 0.8
46 -
A V
5
ISL6531 Functional Pin Description
24 LEAD (SOIC) TOP VIEW
BOOT1 BOOT1 UGATE1 BOOT1 1 2 24 PGND1 23 LGATE1 22 PVCC1 21 OCSET/SD 20 V2_SD 19 PGOOD 18 N/C 17 SENSE2 16 N/C 15 VCC 14 LGATE2 13 PGND2 SENSE1 VREF_IN GNDA GNDA 5 6 7 8 9 PHASE2 10 BOOT2 11 BOOT2 12 UGATE2 13 PGND2 14 PGND2 15 LGATE2 16 VCC 20 N/C 19 SENSE2 18 N/C 17 VCC VREF FB1 COMP1 2 3 4 23 OCSET/SD 22 V2_SD 21 PGOOD PHASE 1 1
32 LEAD 5X5 (QFN) TOP VIEW
UGATE1 UGATE1 LGATE1 26 PGND1 PGND1 PVCC1 25 24 PVCC1
PHASE1 3 VREF 4 FB1 5 COMP1 6
32
31
30
29
28
27
SENSE1 7 VREF_IN 8 GNDA 9
PHASE2 10 BOOT2 11 UGATE2 12
BOOT1 and BOOT2
These pins provide bias voltage to the upper MOSFET drivers. A single capacitor bootstrap circuit may be used to create a BOOT voltage suitable to drive a standard NChannel MOSFET.
(rDS(ON)) set the VDDQ converter overcurrent (OC) trip point according to the following equation:
I OCS * R OCSET I PEAK = ------------------------------------------r DS ( ON )
UGATE1 and UGATE2
Connect UGATE1 and UGATE2 to the corresponding upper MOSFET gate. These pins provide the gate drive for the upper MOSFETs. UGATE2 is also monitored by the adaptive shoot through protection to determine when the upper FET of the VTT regulator has turned off.
An overcurrent trip cycles the soft-start function. Pulling the OCSET/SD pin to ground resets the ISL6531 and all external MOSFETS are turned off allowing the two output voltage power rails to float.
PGOOD
A high level on this open-drain output indicates that both the VDDQ and VTT regulators are within normal operating voltage ranges.
LGATE1 and LGATE2
Connect LGATE1 and LGATE2 to the corresponding lower MOSFET gate. These pins provide the gate drive for the lower MOSFETs. These pins are monitored by the adaptive shoot through protection circuitry to determine when the lower FET has turned off.
GNDA
Signal ground for the IC. Tie this pin to the ground plane through the lowest impedence connection available.
PGND1 and PGND2
These are the power ground connections for the gate drivers of the PWM controllers. Tie these pins to the ground plane through the lowest impedence connection available.
VCC
The 5V bias supply for the chip is connected to this pin. This pin is also the positive supply for the lower gate driver, LGATE2. Connect a well decoupled 5V supply to this pin.
OCSET/SD
A resistor (ROCSET) connected from this pin to the drain of the upper MOSFET of the VDDQ regulator sets the overcurrent trip point. ROCSET, an internal 40A current source (IOCS), and the upper MOSFET on-resistance
V2_SD
A high level on the V2_SD input places the VTT controller into "sleep" mode. In sleep mode, both UGATE2 and LGATE2 are driven low, effectively floating the VTT supply.
6
ISL6531
While the VTT supply "floats", it is held to about 50% of VDDQ via a low current window regulator which drives VTT via the SENSE2 pin. The window regulator can overcome up to at least 10mA of leakage on VTT. While V2_SD is high, PGOOD is low. 300kHz clocks. The clocks are phase locked and displaced 90o to minimize noise coupling between the controllers. The first regulator includes a precision 0.8V reference and is intended to provide the proper VDDQ to a DDRAM memory system. The VDDQ controller implements overcurrent protection utilizing the rDS(ON) of the upper MOSFET. Following a fault condition, the VDDQ regulator is softstarted via a digital soft-start circuit. Included in the ISL6531 is a precision VREF reference -output. VREF is a buffered representation of 1 V DDQ . VREF 2 is derived via a precision internal resistor divider connected to the SENSE1 terminal. The second PWM regulator is designed to provide VTT termination for the DDRAM signal lines. The reference to the VTT regulator is VREF. Thus the VTT regulator provides a -termination voltage equal to 1 V DDQ . The drain of the 2 upper MOSFET of the VTT supply is connected to the regulated VDDQ voltage. The VTT controller is designed to enable both sinking and sourcing current on the VTT rail. Two benefits result from the ISL6531 dual controller -topology. First, as VREF is always 1 V DDQ , the VTT supply 2 will track the VDDQ supply during soft-start cycles. Second, the overcurrent protection incorporated into the VDDQ supply will simultaneously protect the VTT supply.
PHASE1 and PHASE2
Connect PHASE1 and PHASE2 to the corresponding upper MOSFET source. This pin is used as part of the upper MOSFET bootstrapped drives. PHASE1 is used to monitor the voltage drop across the upper MOSFET of the VDDQ regulator for overcurrent protection. The PHASE1 pin is monitored by the adaptive shoot through protection circuitry to determine when the upper FET of the VDDQ supply has turned off.
FB1, COMP1
COMP1 and FB1 are the available external pins of the error amplifier for the VDDQ regulator. The FB1 pin is the inverting inputs of the error amplifier and the COMP1 pin is the associated output. An appropriate AC network across these pins is used to compensate the voltage-controlled feedback loop of the VDDQ converter.
VREF and VREF_IN
VREF produces a voltage equal to one half of the voltage on SENSE1. This low current output is connected to the VREF input of the DDRAM devices being powered. This same voltage is used as the reference input of the VTT error amplifier. Thus VTT is controlled to 50% of VDDQ. VREF_IN is used as an option to overdrive the internal resistor divider network that sets the voltage for both VREF_OUT and the reference voltage for the VTT supply. A 100pF capacitor between VREF_IN and ground is recommended for proper operation.
Initialization
The ISL6531 automatically initializes upon application of input power. Special sequencing of the input supplies is not necessary. The Power-On Reset (POR) function continually monitors the input bias supply voltage at the VCC pin. The POR function initiates soft-start operation after the 5V bias supply voltage exceeds its POR threshold.
Soft-Start
The POR function initiates the digital soft start sequence. The PWM error amplifier reference input for the VDDQ regulator is clamped to a level proportional to the soft-start voltage. As the soft-start voltage slews up, the PWM comparator generates PHASE pulses of increasing width that charge the output capacitor(s). This method provides a rapid and controlled output voltage rise. The soft-start sequence typically takes about 7ms.
-With the VTT regulator reference held at 1 V DDQ it will 2 automatically track the ramp of the VDDQ softstart, thus enabling a soft-start for VTT.
PVCC1
This is the positive supply for the lower gate driver, LGATE1. PVCC1 is connected to a well decoupled 5V.
SENSE1 and SENSE2
Both SENSE1 and SENSE2 are connected directly to the regulated outputs of the VDDQ and VTT supplies, respectively. SENSE1 is used as an input to create the voltage at VREF_OUT and the reference voltage for the VTT supply. SENSE2 is used as the feedback pin of the VTT regulator and as the regulation point for the window regulator that is enabled in V2_SD mode.
Functional Description
Overview
The ISL6531 contains control and drive circuitry for two synchronous buck PWM voltage regulators. Both regulators utilize 5V bootstrapped output topology to allow use of low cost N-Channel MOSFETs. The regulators are driven by
Figure 2 shows the soft-start sequence for a typical application. At T0, the +5V VCC bias voltage starts to ramp. Once the voltage on VCC crosses the POR threshold at time T1, both outputs begin their soft-start sequence. The triangle waveforms from the PWM oscillators are compared to the rising error amplifier output voltage. As the error amplifier voltage increases, the pulse-widths on the UGATE pins increase to reach their steady-state duty cycle at time t2.
7
ISL6531
When the V2_SD input of the ISL6531 is driven high, the VTT regulator is placed into a "sleep" state. In the sleep state the main VTT regulator is disabled, with both the upper and lower MOSFETs being turned off. The VTT bus is maintained -at close to 1 V DDQ via a low current window regulator 2 which drives VTT via the SENSE2 pin. Maintaining VTT at 1 -- V DDQ consumes negligible power and enables rapid 2 wake-up from sleep mode without the need of softstarting the VTT regulator. During this power down mode, PGOOD is held LOW.
VCC (5V) (1V/DIV)
VDDQ (2.5V)
VTT (1.25V) 0V T0 T1 T2 TIME
Output Voltage Selection
The output voltage of the VDDQ regulator can be programmed to any level between VIN (i.e. +5V) and the internal reference, 0.8V. An external resistor divider is used to scale the output voltage relative to the reference voltage and feed it back to the inverting input of the error amplifier, see Figure 3.F However, since the value of R1 affects the values of the rest of the compensation components, it is advisable to keep its value less than 5k. R4 can be calculated based on the following equation:
R1 x 0.8V R4 = ------------------------------------V OUT1 - 0.8V
FIGURE 2. SOFT-START INTERVAL
Shoot-Through Protection
A shoot-through condition occurs when both the upper MOSFET and lower MOSFET are turned on simultaneously, effectively shorting the input voltage to ground. To protect the regulators from a shoot-through condition, the ISL6531 incorporates specialized circuitry which insures that complementary MOSFETs are not ON simultaneously. The adaptive shoot-through protection utilized by the VDDQ regulator looks at the lower gate drive pin, LGATE1, and the phase node, PHASE1, to determine whether a MOSFET is ON or OFF. If PHASE1 is below 0.8V, the upper gate is defined as being OFF. Similarly, if LGATE1 is below 0.8V, the lower MOSFET is defined as being OFF. This method of shoot-through protection allows the VDDQ regulator to source current only. Due to the necessity of sinking current, the VTT regulator employs a modified protection scheme from that of the VDDQ regulator. If the voltage from UGATE2 or from LGATE2 to GND is less than 0.8V, then the respective MOSFET is defined as being OFF and the other MOSFET is turned ON. Since the voltage of the lower MOSFET gates and the upper MOSFET gate of the VTT supply are being measured to determine the state of the MOSFET, the designer is encouraged to consider the repercussions of introducing external components between the gate drivers and their respective MOSFET gates before actually implementing such measures. Doing so may interfere with the shootthrough protection.
If the output voltage desired is 0.8V, simply route VDDQ back to the FB pin through R1, but do not populate R4.
+5V
VCC
D1 BOOT1
C4 UGATE1 ISL6531 PHASE1 LGATE1
Q1
LOUT1
VDDQ
Q2
+ COUT1
FB1 COMP1 C1 R1 C3 C2 R3
R2 R4
FIGURE 3. OUTPUT VOLTAGE SELECTION OF VDDQ
Power Down Mode
DDRAM systems include a sleep state in which the VDDQ voltage to the memories is maintained, but signaling is suspended. During this mode the VTT termination voltage is no longer needed. The only load placed on the VTT bus is the leakage of the associated signal pins of the DDRAM and memory controller ICs. 8
VTT Reference Overdrive
The ISL6531 allows the designer to bypass the internal 50% tracking of VDDQ that is used as the reference for VTT. The ISL6531 was designed to divide down the VDDQ voltage by 50% through two internal matched resistances. These resistances are typically 200k.
ISL6531
One method that may be employed to bypass the internal VTT reference generation is to supply an external reference directly to the VREF_IN pin. When doing this the SENSE1 pin must remain unconnected. Caution must be exercised when using this method as the VTT regulator does not employ a soft start of its own. A second method would be to overdrive the internal resistors. Figure 3 shows how to implement this method. The external resistors used to overdrive the internal resistors should be less than 2k and have a tolerance of 1% or better. This method still supplies a buffer between the resistor network and any loading on the VREF pin. If there is no loading on the VREF pin, then no buffering is necessary and the reference voltage created by the resistor network can be tied directly to VREF.
VDDQ SENSE1 VREF_IN RA VREF + ISL6531
programs the overcurrent trip level (see Figure 1). An internal 40A (typical) current sink develops a voltage across ROCSET that is referenced to VIN. When the voltage across the upper MOSFET of VDDQ (also referenced to VIN) exceeds the voltage across ROCSET , the overcurrent function initiates a soft-start sequence. Figure 5 illustrates the protection feature responding to an overcurrent event on VDDQ. At time t0, an overcurrent condition is sensed across the upper MOSFET of the VDDQ regulator. As a result, both regulators are quickly shutdown and the internal soft-start function begins producing softstart ramps. The delay interval seen by the output is equivalent to three soft-start cycles. The fourth internal softstart cycle initiates a normal soft-start ramp of the output, at time t1. Both outputs are brought back into regulation by time t2, as long as the overcurrent event has cleared. Had the cause of the overcurrent still been present after the delay interval, the overcurrent condition would be sensed and both regulators would be shut down again for another delay interval of three soft start cycles. The resulting hiccup mode style of protection would continue to repeat indefinitely.
RB
VDDQ (2.5V)
TO ERROR AMPLIFIER
VTT (1.25V) FIGURE 4. VTT REFERENCE OVERDRIVE
Converter Shutdown
Pulling and holding the OCSET/SD pin below 0.8V will shutdown both regulators. During this state, PGOOD will be held LOW. Upon release of the OCSET/SD pin, the IC enters into a soft start cycle which brings both outputs back into regulation.
0V
INTERNAL SOFT-START FUNCTION
Voltage Monitoring
The ISL6531 offers a PGOOD signal that will communicate whether the regulation of both VDDQ and VTT are within 15% of regulation, the V2_SD pin is held low and the bias voltage of the IC is above the POR level. If all the criteria above are true, the PGOOD pin will be at a high impedence level. When one or more of the criteria listed above are false, the PGOOD pin will be held low.
DELAY INTERVAL
T0 TIME
T1
T2
FIGURE 5. OVERCURRENT PROTECTION RESPONSE
Overcurrent Protection
The overcurrent function protects the converter from a shorted output by using the upper MOSFET on-resistance, rDS(ON), of VDDQ to monitor the current. This method enhances the converter's efficiency and reduces cost by eliminating a current sensing resistor. The overcurrent function cycles the soft-start function in a hiccup mode to provide fault protection. A resistor (ROCSET) 9
The overcurrent function will trip at a peak inductor current (IPEAK) determined by:
I OCSET x R OCSET I PEAK = ---------------------------------------------------r DS ( ON )
where IOCSET is the internal OCSET current source (40A typical). The OC trip point varies mainly due to the MOSFET rDS(ON) variations. To avoid overcurrent tripping in the
ISL6531
normal operating load range, find the ROCSET resistor from the equation above with: 1. The maximum rDS(ON) at the highest junction temperature. 2. The minimum IOCSET from the specification table. 3. Determine IPEAK for where I is the output inductor ripple current.
) --------I PEAK > I OUT ( MAX ) + ( I - , 2
ISL6531 UGATE1 PHASE1 LGATE1
DDR SDRAM +5V
VDDQ
For an equation for the ripple current see the section under component guidelines titled Output Inductor Selection. A small ceramic capacitor should be placed in parallel with ROCSET to smooth the voltage across ROCSET in the presence of switching noise on the input voltage.
UGATE2 PHASE2 LGATE2
VTT
RT VREF
+ -
FIGURE 6. VTT CURRENT SINKING LOOP
Current Sinking
The ISL6531 VTT regulator incorporates a MOSFET shootthrough protection method which allows the converter to sink current as well as source current. Care should be exercised when designing a converter with the ISL6531 when it is known that the converter may sink current. When the converter is sinking current, it is behaving as a boost converter that is regulating its input voltage. This means that the converter is boosting current into the input rail of the regulator. If there is nowhere for this current to go, such as to other distributed loads on the rail or through a voltage limiting protection device, the capacitance on this rail will absorb the current. This situation will the allow voltage level of the input rail to increase. If the voltage level of the rail is boosted to a level that exceeds the maximum voltage rating of any components attached to the input rail, then those components may experience an irreversible failure or experience stress that may shorten their lifespan. Ensuring that there is a path for the current to flow other than the capacitance on the rail will prevent this failure mode. To insure that the current does not boost up the input rail voltage of the VTT regulator, it is recommended that the input rail of the VTT regulator be the output of the VDDQ regulator. The current being sunk by the VTT regulator will be fed into the VDDQ rail and then drawn into the DDR SDRAM memory module and back into the VTT regulator. Figure 6 shows the recommended configuration and the resulting current loop.
Application Guidelines
Layout Considerations
Layout is very important in high frequency switching converter design. With power devices switching efficiently at 300kHz, the resulting current transitions from one device to another cause voltage spikes across the interconnecting impedances and parasitic circuit elements. These voltage spikes can degrade efficiency, radiate noise into the circuit, and lead to device over-voltage stress. Careful component layout and printed circuit board design minimizes the voltage spikes in the converters. As an example, consider the turn-off transition of the PWM MOSFET. Prior to turn-off, the MOSFET is carrying the full load current. During turn-off, current stops flowing in the MOSFET and is picked up by the lower MOSFET. Any parasitic inductance in the switched current path generates a large voltage spike during the switching interval. Careful component selection, tight layout of the critical components, and short, wide traces minimizes the magnitude of voltage spikes. There are two sets of critical components in a DC-DC converter using the ISL6531. The switching components are the most critical because they switch large amounts o energy, and therefore tend to generate large amounts of noise. Next are the small signal components which connect to sensitive nodes or supply critical bypass current and signal coupling. A multi-layer printed circuit board is recommended. Figure 7 shows the connections of the critical components in the converter. Note that capacitors CIN and COUT could each represent numerous physical capacitors. Dedicate one solid layer, usually a middle layer of the PC board, for a ground plane and make all critical component ground connections with vias to this layer. Dedicate another solid layer as a power plane and break this plane into smaller islands of common voltage levels. Keep the metal runs from the PHASE terminals to the output inductor short. The power plane should support the input power and output power
10
ISL6531
nodes. Use copper filled polygons on the top and bottom circuit layers for the phase nodes. Use the remaining printed circuit layers for small signal wiring. The wiring traces from the GATE pins to the MOSFET gates should be kept short and wide enough to easily handle the 1A of drive current.f
+5V VIN ISL6531 VCC CBP GND BOOT1 CBOOT1 Q1 UGATE1 PHASE1 PHASE1 Q2 LGATE1 PGND1 COMP1 C2A R2A FB1 R4 SENSE1 +5V VIN D2 BOOT2 CBOOT2 Q3 UGATE2 PHASE2 PHASE2 Q4 LGATE2 PGND2 SENSE2 KEY ISLAND ON POWER PLANE LAYER ISLAND ON CIRCUIT PLANE LAYER VIA CONNECTION TO GROUND PLANE COUT2 LOAD LOUT2 VTT VDDQ C1A R1A C3A R3A COUT1 LOAD LOUT1 VDDQ D1 CIN
The critical small signal components include any bypass capacitors, feedback components, and compensation components. Position the bypass capacitor, CBP, close to the VCC pin with a via directly to the ground plane. Place the PWM converter compensation components close to the FB and COMP pins. The feedback resistors for both regulators should also be located as close as possible to the relevant FB pin with vias tied straight to the ground plane as required.
VDDQ Feedback Compensation
This section discusses the feedback compensation of the VDDQ regulator. Figure 8 highlights the voltage-mode control loop for a synchronous-rectified buck converter. The output voltage (VOUT) is regulated to the Reference voltage level. The error amplifier (error amp) output (VE/A) is compared with the oscillator (OSC) triangular wave to provide a pulse-width modulated (PWM) wave with an amplitude of VIN at the PHASE node. The PWM wave is smoothed by the output filter (LO and CO). The modulator transfer function is the small-signal transfer function of VOUT/VE/A . This function is dominated by a DC gain and the output filter (LO and CO), with a double pole break frequency at FLC and a zero at FESR . The DC gain of the modulator is simply the input voltage (VIN) divided by the peak-to-peak oscillator voltage VOSC .
OSC DVOSC DRIVER VIN LO DRIVER PHASE C O ESR (PARASITIC) ZIN REFERENCE VOUT PWM COMPARATOR +
ZFB VE/A + ERROR AMP
DETAILED COMPENSATION COMPONENTS C1 C2 COMP + ISL6531 REFERENCE FIGURE 8. VOLTAGE-MODE BUCK CONVERTER COMPENSATION DESIGN FB R2 ZFB ZIN C3 R1 R3 VOUT
FIGURE 7. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS
The switching components should be placed close to the ISL6531 first. Minimize the length of the connections between the input capacitors, CIN, and the power switches by placing them nearby. Position both the ceramic and bulk input capacitors as close to the upper MOSFET drain as possible. Position the output inductor and output capacitors between the upper MOSFET and lower diode and the load.
Modulator Break Frequency Equations
1 F LC = ----------------------------------------2 x L O x C O 1 F ESR = -----------------------------------------2 x ESR x C O
11
ISL6531
The compensation network consists of the error amplifier (internal to the ISL6531) and the impedance networks ZIN and ZFB. The goal of the compensation network is to provide a closed loop transfer function with the highest 0dB crossing frequency (f0dB) and adequate phase margin. Phase margin is the difference between the closed loop phase at f0dB and 180 degrees. The equations below relate the compensation network's poles, zeros and gain to the components (R1 , R2 , R3 , C1 , C2 , and C3) in Figure 7. Use these guidelines for locating the poles and zeros of the compensation network: 1. Pick gain (R2/R1) for desired converter bandwidth. 2. Place first zero below filter's double pole (~75% FLC). 3. Place second zero at filter's double pole. 4. Place first pole at the ESR zero. 5. Place second pole at half the switching frequency. 6. Check gain against error amplifier's open-loop gain. 7. Estimate phase margin - repeat if necessary. The compensation gain uses external impedance networks ZFB and ZIN to provide a stable, high bandwidth (BW) overall loop. A stable control loop has a gain crossing with -20dB/decade slope and a phase margin greater than 45 degrees. Include worst case component variations when determining phase margin
VTT Feedback Compensation
To ease design and reduce the number of small-signal components required, the VTT regulator is internally compensated. The only stability criteria that needs to be met relates the minimum value of the inductor to the equivalent ESR of the output capacitor bank as shown in the following equation:
L OUT ( MIN ) 20 ( 10
-6
) x ESR OUT x V IN
Compensation Break Frequency Equations
1 F Z1 = --------------------------------2 x R 2 x C 2 1 F P1 = ------------------------------------------------------- C 1 x C 2 2 x R 2 x --------------------- C1 + C2 1 F P2 = ----------------------------------2 x R 3 x C 3
where LOUT(MIN) = minimum output inductor value at full output current ESROUT = equivalent ESR of the output capacitor bank VIN = Input voltage of the converter The design procedure for this output should follow the following steps: 1. Choose the number and type of output capacitors to meet the output transient requirements based on the dynamic loading characteristics of the output. 2. Determine the equivalent ESR of the output capacitor bank and calculate the minimum output inductor value. 3. Verify that the chosen inductor meets this minimum value criteria at full output load. It is recommended that the chosen inductor be no more than 30% saturated at full output load.
1 F Z2 = -----------------------------------------------------2 x ( R 1 + R 3 ) x C 3
Figure 9 shows an asymptotic plot of the DC-DC converter's gain vs frequency. The actual Modulator Gain has a high gain peak due to the high Q factor of the output filter and is not shown in Figure 9. Using the above guidelines should give a Compensation Gain similar to the curve plotted. The open loop error amplifier gain bounds the compensation gain. Check the compensation gain at FP2 with the capabilities of the error amplifier. The Closed Loop Gain is constructed on the graph of Figure 9 by adding the Modulator Gain (in dB) to the Compensation Gain (in dB). This is equivalent to multiplying the modulator transfer function to the compensation transfer function and plotting the gain..
FZ1 100 80 60 GAIN (dB) 40 20 0 -20 -40 -60 R2 20 log ------- R1 MODULATOR GAIN 10 100 FLC 1K FESR 10K 100K 1M LOOP GAIN COMPENSATION GAIN FZ2 FP1 FP2 OPEN LOOP ERROR AMP GAIN
Component Selection Guidelines
Output Capacitor Selection
An output capacitor is required to filter the output and supply the load transient current. The filtering requirements are a function of the switching frequency and the ripple current. The load transient requirements are a function of the slew rate (di/dt) and the magnitude of the transient load current. These requirements are generally met with a mix of capacitors and careful layout. Modern digital ICs can produce high transient load slew rates. High frequency capacitors initially supply the transient and slow the current load rate seen by the bulk capacitors. The bulk filter capacitor values are generally determined by the ESR (effective series resistance) and voltage rating requirements rather than actual capacitance requirements. High frequency decoupling capacitors should be placed as close to the power pins of the load as physically possible. Be careful not to add inductance in the circuit board wiring that
V IN 20 log --------------- V OSC
10M
FREQUENCY (Hz)
FIGURE 9. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN
12
ISL6531
could cancel the usefulness of these low inductance components. Consult with the manufacturer of the load on specific decoupling requirements. Use only specialized low-ESR capacitors intended for switching-regulator applications for the bulk capacitors. The bulk capacitor's ESR will determine the output ripple voltage and the initial voltage drop after a high slew-rate transient. An aluminum electrolytic capacitor's ESR value is related to the case size with lower ESR available in larger case sizes. However, the equivalent series inductance (ESL) of these capacitors increases with case size and can reduce the usefulness of the capacitor to high slew-rate transient loading. Unfortunately, ESL is not a specified parameter. Work with your capacitor supplier and measure the capacitor's impedance with frequency to select a suitable component. In most cases, multiple electrolytic capacitors of small case size perform better than a single large case capacitor. response time to the removal of load. The worst case response time can be either at the application or removal of load. Be sure to check both of these equations at the minimum and maximum output levels for the worst case response time.
Input Capacitor Selection
Use a mix of input bypass capacitors to control the voltage overshoot across the MOSFETs. Use small ceramic capacitors for high frequency decoupling and bulk capacitors to supply the current needed each time Q1 turns on. Place the small ceramic capacitors physically close to the MOSFETs and between the drain of Q1 and the source of Q2 . The important parameters for the bulk input capacitor are the voltage rating and the RMS current rating. For reliable operation, select the bulk capacitor with voltage and current ratings above the maximum input voltage and largest RMS current required by the circuit. The capacitor voltage rating should be at least 1.25 times greater than the maximum input voltage and a voltage rating of 1.5 times is a conservative guideline. The RMS current rating requirement for the input capacitor of a buck regulator is approximately 1/2 the DC load current. The maximum RMS current required by the regulator may be closely approximated through the following equation:
I RMS =
2 V OUT 2 1 V IN - V OUT V OUT --------------- x I OUT + ----- x ------------------------------- x --------------- L x fs V IN V IN 12 MAX
Output Inductor Selection
The output inductor is selected to meet the output voltage ripple requirements and minimize the converter's response time to the load transient. Additionally, the output inductor for the VTT regulator has to meet the minimum value criteria for loop stability as described in the VTT Feedback Compensation section. The inductor value determines the converter's ripple current and the ripple voltage is a function of the ripple current. The ripple voltage and current are approximated by the following equations:
I = VIN - VOUT fs x L x VOUT VIN VOUT = I x ESR
MAX
Increasing the value of inductance reduces the ripple current and voltage. However, the large inductance values reduce the converter's response time to a load transient. One of the parameters limiting the converter's response to a load transient is the time required to change the inductor current. Given a sufficiently fast control loop design, the ISL6531 will provide either 0% or 100% duty cycle in response to a load transient. The response time is the time required to slew the inductor current from an initial current value to the transient current level. During this interval the difference between the inductor current and the transient current level must be supplied by the output capacitor. Minimizing the response time can minimize the output capacitance required. The response time to a transient is different for the application of load and the removal of load. The following equations give the approximate response time interval for application and removal of a transient load:
tRISE = L x ITRAN VIN - VOUT tFALL = L x ITRAN VOUT
For a through hole design, several electrolytic capacitors may be needed. For surface mount designs, solid tantalum capacitors can be used, but caution must be exercised with regard to the capacitor surge currentrating. These capacitors must be capable of handling the surge-current at power-up. Some capacitor series available from reputable manufacturers are surge current tested.
MOSFET Selection/Considerations
The ISL6531 requires two N-Channel power MOSFETs for each PWM regulator. These should be selected based upon rDS(ON) , gate supply requirements, and thermal management requirements. In high-current applications, the MOSFET power dissipation, package selection and heatsink are the dominant design factors. The power dissipation includes two loss components; conduction loss and switching loss. The conduction losses are the largest component of power dissipation for both the upper and the lower MOSFETs. These losses are distributed between the two MOSFETs according to duty factor. The switching losses seen when sourcing current will be different from the switching losses seen when sinking current. The VDDQ regulator will only source current while the VTT regulator can sink and source. When sourcing current, the upper MOSFET realizes most of the switching losses. The lower switch realizes most of the switching losses when the converter is sinking
where: ITRAN is the transient load current step, tRISE is the response time to the application of load, and tFALL is the 13
ISL6531
current (see the equations below). These equations assume linear voltage-current transitions and do not adequately model power loss due the reverse-recovery of the upper and lower MOSFET's body diode. The gate-charge losses are dissipated by the ISL6531 and don't heat the MOSFETs. However, large gate-charge increases the switching interval, tSW which increases the MOSFET switching losses.
LOSSES WHILE SOURCING CURRENT 2 1 P UPPER = Io x r DS ( ON ) x D + -- Io x V IN x t SW x f s 2 PLOWER = Io2 x rDS(ON) x (1 - D) LOSSES WHILE SINKING CURRENT PUPPER = Io2 x rDS(ON) x D 2 1 P LOWER = Io x r DS ( ON ) x ( 1 - D ) + -- Io x V IN x t SW x f s 2 Where: D is the duty cycle = VOUT / VIN , tSW is the combined switch ON and OFF time, and fs is the switching frequency.
GND VCC DBOOT + VD CBOOT UGATEn PHASEn QUPPER NOTE: VG-S VCC -VD QLOWER NOTE: VG-S VCC VIN
BOOTn ISL6531
+
LGATEn
FIGURE 10. UPPER GATE DRIVE BOOTSTRAP
Ensure that both MOSFETs are within their maximum junction temperature at high ambient temperature by calculating the temperature rise according to package thermal-resistance specifications. A separate heat sink may be necessary depending upon MOSFET power, package type, ambient temperature and air flow. Given the reduced available gate bias voltage (5V), logiclevel or sub-logic-level transistors should be used for both NMOSFETs. Caution should be exercised when using devices with very low gate thresholds (VTH). The shoot-through protection circuitry may be circumvented by these MOSFETs. Very high dv/dt transitions on the phase node may cause the Miller capacitance to couple the lower gate with the phase node and cause an undesireable turn on of the lower MOSFET while the upper MOSFET is on.
where QGATE is the maximum total gate charge of the upper MOSFET, CBOOT is the bootstrap capacitance, VBOOT1 is the bootstrap voltage immediately before turn-on, and VBOOT2 is the bootstrap voltage immediately after turn-on. The bootstrap capacitor begins its refresh cycle when the gate drive begins to turn-off the upper MOSFET. A refresh cycle ends when the upper MOSFET is turned on again, which varies depending on the switching frequency and duty cycle. The minimum bootstrap capacitance can be calculated by rearranging the previous equation and solving for CBOOT.
Q GATE C BOOT ---------------------------------------------------V BOOT1 - V
BOOT2
Bootstrap Component Selection
External bootstrap components, a diode and capacitor, are required to provide sufficient gate enhancement to the upper MOSFET. The internal MOSFET gate driver is supplied by the external bootstrap circuitry as shown in Figure 10. The boot capacitor, CBOOT, develops a floating supply voltage referenced to the PHASE pin. This supply is refreshed each cycle, when DBOOT conducts, to a voltage of VCC less the boot diode drop, VD, plus the voltage rise across QLOWER. Just after the PWM switching cycle begins and the charge transfer from the bootstrap capacitor to the gate capacitance is complete, the voltage on the bootstrap capacitor is at its lowest point during the switching cycle. The charge lost on the bootstrap capacitor will be equal to the charge transferred to the equivalent gate-source capacitance of the upper MOSFET as shown:
Q GATE = C BOOT x ( V BOOT1 - V BOOT2 )
Typical gate charge values for MOSFETs considered in these types of applications range from 20 to 100nC. Since the voltage drop across QLOWER is negligible, VBOOT1 is simply VCC - VD. A schottky diode is recommended to minimize the voltage drop across the bootstrap capacitor during the on-time of the upper MOSFET. Initial calculations with VBOOT2 no less than 4V will quickly help narrow the bootstrap capacitor range. For example, consider an upper MOSFET is chosen with a maximum gate charge, Qg, of 100nC. Limiting the voltage drop across the bootstrap capacitor to 1V results in a value of no less than 0.1F. The tolerance of the ceramic capacitor should also be considered when selecting the final bootstrap capacitance value. A fast recovery diode is recommended when selecting a bootstrap diode to reduce the impact of reverse recovery charge loss. Otherwise, the recovery charge, QRR, would have to be added to the gate charge of the MOSFET and taken into consideration when calculating the minimum bootstrap capacitance.
14
ISL6531 ISL6531 DC-DC Converter Application Circuit
Figure 11 shows an application circuit for a DDR SDRAM power supply, including VDDQ (+2.5V) and VTT (+1.25V). Detailed information on the circuit, including a complete Billof-Materials and circuit board description, can be found
+5V
inApplication Note AN9993-of-Materials and circuit board description, can be found in Application Note AN9993 Application Note AN9993-of-Materials and circuit board description, can be found in Application Note AN9993.
R1 3.48k
C1 1000pF
C2 0.1F
D1 C3 1.0F C4,5 150F(x2)
OCSET/SD V2_SD PGOOD VREF
VCC BOOT1 Q1 C6 0.1F
UGATE1
PHASE1
L1 1H Q2
VDDQ @10A C7,8,9,10 150F(x4)
VREF_IN C30 100pF GNDA
PVCC1 LGATE1 C15 0.1F
PGND1 ISL6531 C26 5600pF COMP1 R26 6.34k C27 100pF BOOT2 UGATE2 C16 0.1F PHASE2 R20 1.43k FB1 PGND2 SENSE1 R19 3.01k C25 15000pF SENSE2 LGATE2 D2
C17 1.0F
L2 1H Q3
VTT @5A C18,19 150F(x2)
R25 100
Component Selection Notes:
C4,5,7,8,9,10,18,19 - Each 150mF, Panasonic EEF-UE0J151R D1,2 - Each 30mA Schottky Diode, MA732
L1,2 - Each 1mH Inductor, Panasonic P/N ETQ-P6F1ROSFA Q1,2 - Each Fairchild MOSFET; ITF86130DK8 Q3 - Fairchild MOSFET; ITF86110DK8
FIGURE 11. DDR SDRAM VOLTAGE REGULATOR
15
ISL6531 Small Outline Plastic Packages (SOIC)
N INDEX AREA E -B1 2 3 SEATING PLANE -AD -CA h x 45o H 0.25(0.010) M BM
M24.3 (JEDEC MS-013-AD ISSUE C)
24 LEAD WIDE BODY SMALL OUTLINE PLASTIC PACKAGE INCHES SYMBOL A A1 B MIN 0.0926 0.0040 0.013 0.0091 0.5985 0.2914 MAX 0.1043 0.0118 0.020 0.0125 0.6141 0.2992 MILLIMETERS MIN 2.35 0.10 0.33 0.23 15.20 7.40 MAX 2.65 0.30 0.51 0.32 15.60 7.60 NOTES 9 3 4 5 6 7 8o Rev. 0 12/93
L
C D E e H
0.05 BSC 0.394 0.010 0.016 24 0o 8o 0.419 0.029 0.050
1.27 BSC 10.00 0.25 0.40 24 0o 10.65 0.75 1.27

A1 0.10(0.004) C
h L N
e
B 0.25(0.010) M C AM BS
NOTES: 1. Symbols are defined in the "MO Series Symbol List" in Section 2.2 of Publication Number 95. 2. Dimensioning and tolerancing per ANSI Y14.5M-1982. 3. Dimension "D" does not include mold flash, protrusions or gate burrs. Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006 inch) per side. 4. Dimension "E" does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch) per side. 5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area. 6. "L" is the length of terminal for soldering to a substrate. 7. "N" is the number of terminal positions. 8. Terminal numbers are shown for reference only. 9. The lead width "B", as measured 0.36mm (0.014 inch) or greater above the seating plane, shall not exceed a maximum value of 0.61mm (0.024 inch) 10. Controlling dimension: MILLIMETER. Converted inch dimensions are not necessarily exact.
16
ISL6531 Quad Flat No-Lead Plastic Package (QFN) Micro Lead Frame Plastic Package (MLFP)
L32.5x5
32 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE (COMPLIANT TO JEDEC MO-220VHHD-2 ISSUE C MILLIMETERS SYMBOL A A1 A2 A3 b D D1 D2 E E1 E2 e k L L1 N Nd Ne P 8 0.25 0.30 2.95 2.95 0.18 MIN 0.80 NOMINAL 0.90 0.20 REF 0.23 5.00 BSC 4.75 BSC 3.10 5.00 BSC 4.75 BSC 3.10 0.50 BSC 0.40 32 8 8 0.60 12 0.50 0.15 3.25 3.25 0.30 MAX 1.00 0.05 1.00 NOTES 9 9 5,8 9 7,8 9 7,8 8 10 2 3 3 9 9 Rev. 1 10/02 NOTES: 1. Dimensioning and tolerancing conform to ASME Y14.5-1994. 2. N is the number of terminals. 3. Nd and Ne refer to the number of terminals on each D and E. 4. All dimensions are in millimeters. Angles are in degrees. 5. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 identifier may be either a mold or mark feature. 7. Dimensions D2 and E2 are for the exposed pads which provide improved electrical and thermal performance. 8. Nominal dimensions are provided to assist with PCB Land Pattern Design efforts, see Intersil Technical Brief TB389. 9. Features and dimensions A2, A3, D1, E1, P & are present when Anvil singulation method is used and not present for saw singulation. 10. Depending on the method of lead termination at the edge of the package, a maximum 0.15mm pull back (L1) maybe present. L minus L1 to be equal to or greater than 0.3mm.
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Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com 17

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